Control scheme for distortion reduction

ABSTRACT

In predistorter ( 100 ), non-linear element ( 126 ) generates a distortion signal which is injected into the input ( 110 ) of a non-linear power amplifier ( 112  at  118 ). The distortion signal is optimised by a vector modulator to cancel distortion caused by amplifier ( 112 ). The control signals for the vector modulator are provided by a digital signal processor (FIG.  2 ). The input signal ( 110 ) and the output signal ( 114 ) are sampled and processed to produce an audio frequency error signal which is correlated in the DSP to produce the control signals for the vector modulator whilst avoiding analogue correlation problems.

This invention relates to a method of, and apparatus for, producingcontrol signals to control a distortion reducing mechanism, such as maybe used to linearise the output of a non-linear power amplifier.

A known control scheme for an amplifier predistortion arrangementcomprises, as described in WO 99/45640, a mixer (e.g. 685 in thatdocument) for multiplying a reference signal with a feedback signalderived from the output of the amplifier, which latter signal containsresidual distortion. The mixing process is equivalent to a correlationprocess in that any component in the feedback signal which is equivalentin frequency to a component in the reference signal is mixed down tobaseband and contributes to the DC component (i.e. at 0 Hz) of the mixeroutput. The mixer output is then integrated to remove AC componentstherein and is then used as a feedback control signal for thepredistorter. That is, the DC component is isolated as the wantedsignal. In the mixer, the type of detection performed is known ascoherent detection since the wanted signal always appears at the samefrequency, 0 Hz (unless, that is, the feedback signal supplied to themixer has been frequency shifted by a local oscillator (LO), in whichcase the wanted signal always appears at the LO frequency.) This isdistinct from incoherent conversion, where the wanted signal componentwill not always be mixed down to the same frequency.

In the control scheme described above, there are problems will DCoffsets and offset drift in the correlation process. This is due to thecorrelation processes being performed in the analogue domain, resultingin a DC output. Ideally, this DC level would be directly proportional tothe quantity being measured, (e.g. the amount of residualintermodulation distortion present), and would result in a zero voltoutput when the quantity has been minimised, (i.e. eliminated).

Unfortunately, analogue correlators (e.g. mixers or multipliers) moretypically have a DC offset and hence their output will not fall to zerowhen the correlation result is minimised. In addition, this offset valuewill drift with time, temperature changes and input signal levelchanges, and hence it is usually difficult to use subtraction to cancelthe offset with any degree of accuracy. The presence of such offsets andtheir fluctuating nature limits the achievable linearisation performanceof the predistorter.

According to a first aspect, the invention provides a method of reducingthe appearance of distortion in an output signal which a signal handlingmeans produces in response to an input signal, the method comprising:sampling both the input and output signals; frequency shifting one ofthe sampled signals by a frequency offset amount; converting, bydetection, the frequency-offset sampled signal and the other sampledsignal to baseband signals, processing the baseband signals to producecontrol signals; and predistorting the input signal under control of thecontrol signals.

According to a second aspect, the invention also provides apparatus forreducing the appearance of distortion in an output signal which a signalhandling means produces in response to an input signal, the apparatuscomprising means for sampling both the input and output signals, meansfor frequency shifting one of the sampled signals by a frequency offsetamount, means for converting, by detection, the frequency offset sampledsignal and the other sampled signal to baseband signals, means forprocessing the baseband signals to produce control signals, and meansfor predistorting the input signal under control of the control signals.

The invention advantageously uses detector-based down-conversion, ratherthan local oscillator based down conversion with its associateddisadvantages. In the present context, frequency conversion to basebandby detection relates to frequency conversion to baseband without the useof local oscillator signals. It can either involve implicitmultiplication (by, e.g. square-law detectors which output the square oftheir input), or can be achieved by correlation of, for example, aninput signal with a related output signal (using, e.g. a mixer, amultiplier, or a non-linear device such as a diode). No additional orauxiliary signals (such as local oscillators and the like) are requiredto perform the frequency conversion.

Another advantage of the invention is that it produces baseband signalssuitable for conversion to digital signals to allow the control signalsto be produced in the digital domain in, for example, a digital signalprocessor. This means that the process of producing the control signalsmay be performed using digital correlation thus avoiding DC-offsets andDC-offset drift associated with mixers and multipliers used in analoguecorrelation processors.

In a preferred embodiment, the sampled input signal is multiplied withitself to produce a first reference signal which is one order higherthan a target distortion component of a particular order in the outputsignal. The sampled input signal may be multiplied with the sampledoutput signal to produce a second reference signal. The first and secondreference signals may be multiplied together to produce a thirdreference signal at the offset frequency. The third reference signal maybe multiplied with a signal at the offset frequency in the digitaldomain to produce DC signals for controlling the predistortion process.

In an alternative embodiment, however, the third reference signal may bemultiplied with a signal at the offset frequency on the analogue domain.

As mentioned above, producing the DC control signals in the digitaldomain may substantially eliminate the problems of DC offset and DCoffset drift associated with analogue methods of producing controlsignals. The target distortion component may be a third order distortioncomponent, and the control signals developed by correlating with thetarget distortion component may be used to control the predistorter inthe suppression of the target distortion component or a wider spectrumof distortion.

The preferred embodiment may also include multiplying the sampled inputsignal with itself to produce further reference signals, each of whichis one order higher than a corresponding target distortion component ofa specific order appearing in the output signal, and multiplying eachfurther reference signal with the second reference signal to producemodified further reference signals at the offset frequency. The modifiedfurther reference signals may be multiplied with a signal at the offsetfrequency to produce DC control signals for controlling thepredistortion process to substantially eliminate distortion appearing inthe output signal and corresponding to the respective target distortioncomponents. In this way, the system can be extended to individualcontrol of the predistorter to combat distortion appearing at individualtarget distortion components.

In a preferred embodiment, the predistortion process is diode, FET,Bipolar transistor, dual-gate FET or mixer based.

By way of example only, certain embodiments of the invention will now bedescribed with reference to the accompanying drawings, in which:

FIG. 1 is a schematic diagram of a predistorter arrangement forlinearising an amplifier;

FIG. 2 is a schematic diagram of a digital signal processor operating inthe system of FIG. 1;

FIG. 3 is a second predistorter arrangement for linearising anamplifier; and

FIG. 4 is a third predistorter arrangement for linearising an amplifier.

The predistorter 100 shown in FIG. 1 operates on the RF input signal 110to non-linear power amplifier 112 in order to reduce distortionappearing in its output 114.

The predistorter 100 comprises a splitter 116 which removes a portion ofthe RF input 110 to form the basis of a predistortion signal which isrecombined with the input signal at combiner 118. The portion of theinput signal removed at splitter 116 is adjusted to an appropriateamplitude level by automatic level controller 120. The desired amplitudelevel is set by a control signal from a digital signal processor (shownin FIG. 2). The extracted portion of the RF input signal is thenamplified at 122 and supplied to splitter 124. The input signal issupplied from splitter 124 to non-linear element 126 which produces athird order component of the input signal. The third order component isconverted into inphase and quadrature components at 128. Each of theinphase and quadrature components is multiplied (at 130 and 132) withrespective control values produced by a digital signal processor (shownin FIG. 2). The modulated inphase and quadrature components are thencombined at 134 to produce a predistortion signal. The predistortionsignal is amplified at 136 and injected into the main RF input signalpath at 118. The predistorter 100 can be switched on and off byswitching amplifier 136 on and off. This allows the predistorter 100 tobe disabled at low input signal levels where distortion is notintolerable and where predistortion may contribute to, rather thansuppress, distortion appearing in the output 114. Alternatively, theon/off function can be provided by means of a PIN diode switch, or otherform of RF switch (e.g. FET, relay, etc.). A time delay 138 is providedin the main RF input signal path so that the signals arriving atcombiner 118 are appropriately time aligned.

Splitter 124 also provides the sampled RF input signal to a furthersplitter 140 via a time delay 142 and an amplifier 144. The signals fromsplitter 140 are processed in conjunction with a portion of the outputsignal of amplifier 112 removed at splitter 146. Time delay 142appropriately time-aligns the signal provided to splitter 140 with thesignal fed back from splitter 146. The sampled input signal is providedfrom splitter 140 to both inputs of a mixer 148 which outputs a squaredversion of the input signal. The squared signal produced by mixer 148 islow pass filtered at 150, amplified at 152, filtered again at 154, andsquared again using mixer 156 to produce a fourth order signal. Wherethe input 110 comprises a two tone test signal, then the fourth ordersignal will be a fourth order signal at baseband, that is, a signal witha frequency twice that of the tone spacing in a two-tone test (withvirtually no leakage at the tone spacing frequency). The fourth ordersignal produced by mixer 156 contains a DC component which varies withthe input signal level. This DC component is extracted by low passfilter 158 and supplied to the DSP digital signal processor (shown inFIG. 2) to produce an appropriate control signal for the automatic levelcontroller 120.

The portion of the RF output 114 removed at splitter 146 is attenuatedat 160. The attenuated amplifier output sample is frequency offset by asmall amount using an audio frequency tone 162 and an image reject mixer164. The audio frequency tone 162 is supplied by the DSP to facilitatesubsequent correlation. The frequency offset amplifier output sample isthen amplified at 166 and supplied to mixer 168. The other input ofmixer 168 is supplied with the sampled amplifier input signal, providedby splitter 140 (after being subject to time delay 142 for timealignment purposes). The mixing process at 168 creates a basebandspectrum.

If one considers, for example, a case where amplifier 112 exhibits onlythird order non-linearity, then, where the input 110 comprises a twotone test signal, the baseband spectrum consists of two frequencycomponents. The two components are a component at the tone differencefrequency, offset by an amount equal to the frequency of the injectedaudio frequency signals 162, and a component at double the tone spacing,again offset by an amount equal to the frequency of the injected audiofrequency signals 162. It is the latter component which is of interestin the control system, as this component contains information about thelevel of the third order intermodulation present in the spectrum of theamplifier output 114, without corruption from the main signal energy ofthe (downconverted) input signals. The former component also containsthis information, but it is masked by the main signal energy of the(downconverted) input signals. The output of mixer 168 is then low passfiltered at 170 and subjected to amplification at 172. Any audiofrequency tone feed through component in the baseband signals producedby mixer 168 is blocked by audio frequency tone reject filter 174 priorto the baseband signals undergoing correlation processing.

At mixer 176, the frequency offset, downconverted output of amplifier112 is correlated with the fourth order reference signal produced bysquaring mixer 156. The resulting output of mixer 176 is an error toneat the audio offset frequency, and it contains the gain and phaseinformation needed to steer the predistorter 100 to optimum performance.This signal is isolated by bandpass filter 178 and fed to the DSP.

The DSP, which also forms a part of predistorter 100, is shown in FIG.2. The DSP implements an audio frequency oscillator 200 which providesinphase and quadarature versions of an audio frequency signal, 210 and212 respectively, for use in the rest of the system. The signals 210 and212 are converted into analogue signals and supplied on lines 162 toimage reject mixer 164 to frequency offset the sampled amplifier output114. The signals 210 and 212 are also fed to respective correlatingmixers 214 and 216. The remaining input of each of mixers 214 and 216 issupplied with the audio frequency error signal provided by filter 178(after appropriate analogue to digital conversion). The outputs ofcorrelating mixers 214 and 216 are supplied to vector control matrix 218which supplies weighted sums of the correlator outputs to integrators220 and 222. The weighting factors are determined by attenuators 224 to230 in vector control matrix 218. The vector control matrix allowscontrol system phase offset to be adjusted and eliminated if necessary.The outputs of integrators 220 and 222 provide, respectively, the I andQ control inputs for the mixers 130 and 132 in the vector modulatorwhich controls the amplitude and phase characteristics of thepredistortion which is injected into the input signal for amplifier 112.The DSP also receives an automatic level control output from mixer 158(again, after appropriate analogue to digital conversion). This signalis supplied to integrator 232 which has an appropriate offset value. Thefunction of the integrator 232 is to steer the automatic level controlsignal to be equal to the offset value. The offset value is chosen suchthat the drive level of the non-linear element is optimised. The signaloutput by integrator 232 is converted into an analogue control signaland supplied to automatic level controller 120 in FIG. 1.

An alternative version of the predistorter 100 described in FIG. 1 isshown in FIG. 3. In FIG. 3, components carried over from FIG. 1 retainthe same reference numerals and their functions will not be describedagain in detail. In the predistorter 300 of FIG. 3, it is the sampledinput signals 110 which are frequency offset by image reject mixing withthe audio frequency tone provided by the DSP, instead of the sampledoutput signal 114. The image reject mixer 310 is located between timedelay 142 and splitter 140. The operation of predistorter 300 is broadlythe same as that of predistorter 100 and the DSP used in conjunctionwith predistorter 300 may perform in the same way as the DSP used inconjunction with predistorter 100. The main difference is thatcorrelation may be performed at three times the audio tone frequency asan alternative to correlating at the tone frequency itself. This mayease the tone feed through specification/filtering in some applications.

The predistorter 400, shown in FIG. 4, is a modified version of thepredistorter 100, shown in FIG. 1. To simplify the description, theautomatic level control process (120 in FIG. 1) has been omitted and thetwo consecutive squaring processes used to produce the fourth ordersignal used in controlling the third order predistortion component havebeen abbreviated as a single fourth order generation process 430.

The predistorter 400 includes a fifth order non-linear element 410 forgenerating a fifth order component from the RF input signal. The fifthorder component is adjusted by vector modulator 412 under the control ofsignals 414 from the DSP and is then injected into the input signal toamplifier 416 at combiner 418. The vector modulator 412 adjusts thefifth order signal to cancel fifth order IMD in the output of amplifier416. The process for generating the control signals 414 will now bedescribed.

A portion of the time delayed, sampled RF input signal, is removed atsplitter 420 and is supplied to process 422 which produces a sixth ordercomponent from the input signal. The output of mixer 424 is correlatedwith the sixth order signal at mixer 426 to produce an error signal 428which is used by the DSP to generate the control signals 414. The DSPhandles the signal 428 in an analogous manner to the error signal 178 inFIG. 1. The error signal 428 is mixed with inphase and quadratureversions of a local oscillator signal at the offset frequency to produceI and Q signals for the vector modulator 412. The control scheme steersthe third and fifth order predistortion components to minimise the thirdand fifth order IMD distortion appearing in the sampled output signal ofamplifier 416.

It will be apparent that this system can be extended to further, higherorder predistortion components (for example, seventh, ninth, etc.components) in a straightforward manner. Additionally, it is possible tomix the audio frequency signal into the sampled input signal rather thanthe sampled output signal, in the manner shown in FIG. 3.

What is claimed is:
 1. Apparatus for reducing the appearance ofdistortion in an output signal which signal handling equipment producesin response to an input signal, the apparatus comprising: a sampler forsampling both the input and output signals; a shifter for frequencyshifting one of the sampled signals by a frequency offset amount; aconverter for converting, by incoherent detection, the frequency-offsetsampled signal and the other sampled signal to baseband signals, acontroller for processing the baseband signals to produce controlsignals; and a predistorter for predistorting the input signal undercontrol of the control signals.
 2. Apparatus according to claim 1,including a multiplier for multiplying the sampled input signal withitself to produce a first reference signal which is one order higherthan a target distortion component of a particular order in the outputsignal.
 3. Apparatus according to claim 2, including a multiplier formultiplying the sampled input signal with the sampled output signal toproduce a second reference signal.
 4. Apparatus according to claim 3,including a multiplier for multiplying the first and second referencesignals together to produce a third reference signal at the offsetfrequencey.
 5. Apparatus according to claim 4, including a multiplierfor multiplying, in the digital or analog domain, the third referencesignal with a signal at the offset frequency to produce DC signals forcontrolling the predistorter.
 6. Apparatus according to claim 2, whereinthe target distortion component is a third order distortion component.7. Apparatus according to claim 1, wherein the predistorter reducesdistortion in the form of one specific distortion component of onespecific order.
 8. Apparatus according to claim 1, wherein thepredistorter reduces distortion in the form of several distortioncomponents of different orders.
 9. Apparatus according to claim 8,further comprising a multiplier for multiplying the sampled input signalwith itself to produce further reference signals each one order higherthan a corresponding target distortion component appearing in the outputsignal.
 10. Apparatus acoording to claim 9, further comprising amultiplier for multiplying each further reference signal with the secondreference signal to produce modified further reference signals at theoffset frequency.
 11. Apparatus according to claim 10, comprising amultiplier for multiplying, in the digital or analog domain, eachmodified further reference signal with a signal at the offset frequencyto produce DC signals for controlling the predistorter.
 12. Apparatusaccording to claim 1, wherein the sampled input signal is the one of thesampled signals which is frequency shifted by the frequency offsetamount.
 13. Apparatus according to claim 1, wherein the sampled outputsignal is the one of the sampled signals which is frequency shifted bythe frequency offset amount.
 14. Apparatus according to claim 1, whereinthe DC signals are used to control the I and Q paths of one or morevector modulators in the predistorter.
 15. Apparatus according to claim1, wherein the DC signals are used for controlling amplitude and/orphase of one or more signals in the predistorter.
 16. Apparatusaccording to claim 1, wherein the signal handling equipment comprisesone or more amplifiers.
 17. Apparatus according to claim 1, wherein thepredistorter comprises a multiplier for multiplying the input signalwith itself to produce a predistortion signal for injection into theinput signal.
 18. A method of reducing the appearance of distortion inan output signal which signal handling equipment produces in response toan input signal, the method comprising: sampling both input and outputsignals; frequency shifting one of the sampled signals by a frequencyoffset amount; converting, by incoherent detection, the frequency-offsetsampled signal and the other sampled signal to baseband signals.processing the baseband signals to produce control signals; andpredistorting the input signal under control of the control signals.